Systems and methods for high precision and/or low loss regulation of output currents of power conversion systems

ABSTRACT

Systems and methods are provided for signal processing. An example error amplifier for processing a reference signal and an input signal associated with a current of a power conversion system includes a first operational amplifier, a second operational amplifier, a first transistor, a second transistor, a current mirror component, a switch, a first resistor and a second resistor. The first operational amplifier includes a first input terminal, a second input terminal and a first output terminal, the first input terminal being configured to receive a reference signal. The first transistor includes a first transistor terminal, a second transistor terminal and a third transistor terminal, the first transistor terminal being configured to receive a first amplified signal from the first output terminal, the third transistor terminal being coupled to the second input terminal.

1. CROSS-REFERENCES TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.13/969,281, filed Aug. 16, 2013, which claims priority to Chinese PatentApplication No. 201310306106.X, filed Jul. 19, 2013, both of theseapplications being commonly assigned and incorporated by referenceherein for all purposes.

2. BACKGROUND OF THE INVENTION

The present invention is directed to integrated circuits. Moreparticularly, the invention provides a system and method for currentregulation. Merely by way of example, the invention has been applied topower conversion systems. But it would be recognized that the inventionhas a much broader range of applicability.

Light emitting diodes (LEDs) are widely used for lighting applications.Oftentimes, the currents flowing through LEDs need to be approximatelyconstant. The output-current precision of LEDs is usually used fordetermining the constant-current properties of a LED lighting system.

FIG. 1 is a simplified diagram showing a conventional power conversationsystem for LED lighting. The power conversion system 100 includes acontroller 102, resistors 104, 124, 126 and 132, capacitors 106, 120 and134, a diode 108, a transformer 110 including a primary winding 112, asecondary winding 114 and an auxiliary winding 116, a power switch 128,a current sensing resistor 130, and a rectifying diode 118. Thecontroller 102 includes terminals 138, 140, 142, 144, 146 and 148. Forexample, the power switch 128 is a bipolar junction transistor. Inanother example, the power switch 128 is a MOS transistor.

An alternate-current (AC) input voltage 152 is applied to the system100. A bulk voltage 150 (e.g., a rectified voltage no smaller than 0 V)associated with the AC input voltage 152 is received by the resistor104. The capacitor 106 is charged in response to the bulk voltage 150,and a voltage 154 is provided to the controller 102 at the terminal 138(e.g., terminal VCC). If the voltage 154 is larger than a predeterminedthreshold voltage (e.g., a under-voltage lock-out threshold) inmagnitude, the controller 102 begins to operate normally, and outputs adriving signal 156 through the terminal 142 (e.g., terminal GATE). Forexample, the driving signal 156 is a pulse-width-modulation (PWM) signalwith a switching frequency and a duty cycle. The switch 128 is closed(e.g., being turned on) or open (e.g., being turned off) in response tothe driving signal 156 so that the output current 158 is regulated to beapproximately constant.

The auxiliary winding 116 charges the capacitor 106 through the diode108 when the switch 128 is closed (e.g., being turned on) in response tothe driving signal 156 so that the controller 102 can operate normally.A feedback signal 160 is provided to the controller 102 through theterminal 140 (e.g., terminal FB) in order to detect the ending of ademagnetization process of the secondary winding 118 for charging ordischarging the capacitor 134 using an internal error amplifier in thecontroller 102. The resistor 130 is used for detecting a primary current162 flowing through the primary winding 112, and a current-sensingsignal 164 is provided to the controller 102 through the terminal 144(e.g., terminal CS) to be processed during each switching cycle. Peakmagnitudes of the current-sensing signal 164 are sampled and provided tothe internal error amplifier. The capacitor 120 is used to keep anoutput voltage 168 stable.

FIG. 2 is a simplified conventional diagram showing the controller 102as part of the system 100. The controller 102 includes an oscillator202, an under-voltage lock-out (UVLO) component 204, a modulationcomponent 206, a logic controller 208, a driving component 210, ademagnetization detector 212, an error amplifier 216, and acurrent-sensing component 214.

As shown in FIG. 2, the UVLO component 204 detects the signal 154 andoutputs a signal 218. If the signal 154 is larger than a predeterminedthreshold in magnitude, the controller 102 begins to operate normally.If the signal 154 is smaller than the predetermined threshold inmagnitude, the controller 102 is turned off. The error amplifier 216receives a signal 220 from the current-sensing component 214 and areference signal 222 and outputs an amplified signal 224 to themodulation component 206. The modulation component 206 also receives asignal 228 from the oscillator 202 and outputs a modulation signal 226which is a PWM signal. For example, the signal 228 is a ramping signaland increases, linearly or non-linearly, to a peak magnitude during eachswitching period. In another example, the modulation signal 226 has afixed switching frequency and the duty cycle of the signal 226 isdetermined based on a comparison between the signal 224 and the signal228. The logic controller 208 processes the modulation signal 226 andoutputs a control signal 230 to the driving component 210 whichgenerates the signal 156 to turn on or off the switch 128. Thedemagnetization detector 212 detects the feedback signal 160 and outputsa signal 232 for determining the beginning and the end of thedemagnetization process of the secondary winding 114.

FIG. 3 is a simplified conventional diagram showing the current-sensingcomponent 214 and the error amplifier 216 as parts of the controller102. The current-sensing component 214 includes a switch 302 and acapacitor 304. The error amplifier 216 includes switches 306 and 308, anoperational amplifier 310.

As shown in FIG. 3, the current-sensing component 214 samples thecurrent-sensing signal 164 and the error amplifier 216 amplifies thedifference between the signal 220 and the reference signal 222.Specifically, the switch 302 is closed (e.g., being turned on) or open(e.g., being turned off) in response to a signal 314 in order to samplepeak magnitudes of the current-sensing signal 164 in different switchingperiods. If the switch 302 is closed (e.g., being turned on) in responseto the signal 314 and the switch 306 is open (e.g., being turned off) inresponse to the signal 232 from the demagnetization detector 212, thecapacitor 304 is charged and the signal 220 increases in magnitude. Ifthe switch 306 is closed (e.g., being turned on) in response to thesignal 232, the switch 308 is open (e.g., being turned off) in responseto a signal 312 and the difference between the signal 220 and thereference signal 222 is amplified by the amplifier 310. For example,during the demagnetization process of the secondary winding 114, thesignal 232 is at a logic high level. The switch 306 remains closed(e.g., being turned on) and the switch 308 remains open (e.g., beingturned off). The amplifier 310, together with the capacitor 134,performs integration associated with the signal 220.

Under stable normal operations, an average output current is determined,according to the following equation, without taking into account anyerror current:

$\begin{matrix}{\overset{\_}{I_{o}} = {\frac{1}{2} \times N \times \frac{V_{ref\_ ea}}{R_{cs}}}} & \left( {{Equation}\mspace{14mu} 1} \right)\end{matrix}$where N represents a turns ratio between the primary winding 112 and thesecondary winding 114, V_(ref) _(_) _(ea) represents the referencesignal 222 and R_(cs) represents the resistance of the resistor 130.

But the system 100 has problems in regulating the output current to beapproximately constant. Hence it is highly desirable to improve thetechniques of regulating output currents of power conversion systems.

3. BRIEF SUMMARY OF THE INVENTION

The present invention is directed to integrated circuits. Moreparticularly, the invention provides a system and method for currentregulation. Merely by way of example, the invention has been applied topower conversion systems. But it would be recognized that the inventionhas a much broader range of applicability.

According to one embodiment, an error amplifier for processing areference signal and an input signal associated with a current of apower conversion system includes a first operational amplifier, a secondoperational amplifier, a first transistor, a second transistor, acurrent mirror component, a switch, a first resistor and a secondresistor. The first operational amplifier includes a first inputterminal, a second input terminal and a first output terminal, the firstinput terminal being configured to receive a reference signal. The firsttransistor includes a first transistor terminal, a second transistorterminal and a third transistor terminal, the first transistor terminalbeing configured to receive a first amplified signal from the firstoutput terminal, the third transistor terminal being coupled to thesecond input terminal. The second operational amplifier includes a thirdinput terminal, a fourth input terminal and a second output terminal,the third input terminal being configured to receive an input signalassociated with a current flowing through a primary winding of a powerconversion system. The second transistor includes a fourth transistorterminal, a fifth transistor terminal and a sixth transistor terminal,the fourth transistor terminal being configured to receive a secondamplified signal from the second output terminal, the sixth transistorterminal being coupled to the fourth input terminal. The current mirrorcomponent includes a first component terminal and a second componentterminal, the first component terminal being coupled to the secondtransistor terminal. The switch includes a first switch terminal and asecond switch terminal, the first switch terminal being coupled to thesecond component terminal, the second switch terminal being coupled tothe fifth transistor terminal. The first resistor includes a firstresistor terminal and a second resistor terminal, the first resistorbeing associated with a first resistance, the first resistor terminalbeing coupled to the second input terminal. The second resistor includesa third resistor terminal and a fourth resistor terminal, the secondresistor being associated with a second resistance, the third resistorterminal being coupled to the fourth input terminal. The firstresistance is larger than the second resistance in magnitude. The secondcomponent terminal is configured to output an output signal based on atleast information associated with the reference signal and the inputsignal.

According to another embodiment, a system controller for regulating apower conversion system includes a first controller terminal configuredto receive a first voltage associated with a first current, the firstcurrent being related to an input voltage of a power conversion system,a compensation component coupled to the first controller terminal andconfigured to, if the first voltage satisfies one or more firstconditions, generate a compensation current based on at leastinformation associated with the first current, and a second controllerterminal coupled to the compensation component and configured to providea compensation voltage based on at least information associated with thecompensation current, the compensation voltage being equal in magnitudeto the compensation current multiplied by a compensation resistance, thecompensation resistance being associated with a compensation resistor.The system controller further includes a current sensing componentconfigured to receive a second voltage and generate an output signal,the second voltage being equal to a sum of a third voltage and thecompensation voltage in magnitude, the third voltage being proportionalto a second current flowing through a primary winding of the powerconversion system, and an error amplifier configured to receive theoutput signal and a reference signal, generate an amplified signal basedon at least information associated with the output signal and thereference signal, and output the amplified signal to affect a switchassociated with the second current.

According to yet another embodiment, a system controller for regulatinga power conversion system includes a first controller terminalconfigured to receive a feedback signal associated with an outputvoltage of a power conversion system, a compensation component coupledto the first controller terminal and configured to sample the feedbacksignal during a demagnetization process of the power conversion systemand generate a compensation current based on at least informationassociated with the sampled feedback signal, and a second controllerterminal coupled to the compensation component and configured to providea compensation voltage based on at least information associated with thecompensation current, the compensation voltage being equal in magnitudeto the compensation current multiplied by a compensation resistance, thecompensation resistance being associated with a compensation resistor.In addition, the system controller includes a current sensing componentconfigured to receive an input voltage and generate an output signal,the input voltage being equal to a sum of a first voltage and thecompensation voltage in magnitude, the first voltage being proportionalto a first current flowing through a primary winding of the powerconversion system, and an error amplifier configured to receive theoutput signal and a reference signal, generate an amplified signal basedon at least information associated with the output signal and thereference signal, and output the amplified signal to affect a switchassociated with the first current.

According to yet another embodiment, a system controller for regulatinga power conversion system includes a first controller terminalconfigured to receive a feedback signal associated with an outputvoltage of a power conversion system, a compensation component coupledto the first controller terminal and configured to sample the feedbacksignal during a demagnetization process of the power conversion systemand generate a compensation current based on at least informationassociated with the sampled feedback signal, and an error amplifierincluding a first input terminal, a second input terminal, and an outputterminal coupled to the compensation component. The first input terminalis configured to receive an input voltage, the second input terminal isconfigured to receive a reference voltage, and the output terminal isconfigured to output a first output current related to a differencebetween the input voltage and the reference voltage in magnitude. Theerror amplifier and the compensation component are further configured togenerate a second output current equal to a difference between the firstoutput current and the compensation current in magnitude.

In one embodiment, a method for regulating a power conversion systemincludes receiving a first voltage associated with a first current, thefirst current being related to an input voltage of a power conversionsystem, generating, if the first voltage satisfies one or more firstconditions, a compensation current based on at least informationassociated with the first current, and providing an compensation voltagebased on at least information associated with the compensation current,the compensation voltage being equal in magnitude to the compensationcurrent multiplied by a compensation resistance, the compensationresistance being associated with a compensation resistor. The methodfurther includes receiving an input voltage, the input voltage beingequal to a sum of a second voltage and the compensation voltage inmagnitude, the second voltage being proportional to a second currentflowing through a primary winding of the power conversion system,generating an output signal based on at least information associatedwith the input voltage, and receiving the output signal and a referencesignal. In addition, the method includes generating an amplified signalbased on at least information associated with the output signal and thereference signal, and outputting the amplified signal in order to affecta switch associated with the second current.

In another embodiment, a method for regulating a power conversion systemincludes receiving a feedback signal associated with an output voltageof a power conversion system, sampling the feedback signal during ademagnetization process of the power conversion system, and generating acompensation current based on at least information associated with thesampled feedback signal. The method further includes providing acompensation voltage based on at least information associated with thecompensation current, the compensation voltage being equal in magnitudeto the compensation current multiplied by a compensation resistance, thecompensation resistance being associated with a compensation resistor,receiving an input voltage, the input voltage being equal to a sum of afirst voltage and the compensation voltage in magnitude, the firstvoltage being proportional to a first current flowing through a primarywinding of the power conversion system, and generating an output signal.In addition, the method includes receiving the output signal and areference signal, generating an amplified signal based on at leastinformation associated with the output signal and the reference signal,and outputting the amplified signal to affect a switch associated withthe first current.

In yet another embodiment, a method for regulating a power conversionsystem includes receiving a feedback signal associated with an outputvoltage of a power conversion system, sampling the feedback signalduring a demagnetization process of the power conversion system, andgenerating a compensation current based on at least informationassociated with the sampled feedback signal. Additionally, the methodincludes receiving an input voltage at a first input terminal of anerror amplifier, and the error amplifier further includes a second inputterminal and an output terminal. Moreover, the method includes receivinga reference voltage at the second input terminal, generating a firstoutput current at the output terminal related to a difference betweenthe input voltage and the reference voltage in magnitude, and outputtinga second output current equal to a difference between the first outputcurrent and the compensation current in magnitude.

Depending upon embodiment, one or more benefits may be achieved. Thesebenefits and various additional objects, features and advantages of thepresent invention can be fully appreciated with reference to thedetailed description and accompanying drawings that follow.

4. BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified diagram showing a conventional power conversationsystem for LED lighting.

FIG. 2 is a simplified conventional diagram showing the controller aspart of the system as shown in FIG. 1.

FIG. 3 is a simplified conventional diagram showing the current-sensingcomponent and the error amplifier as parts of the controller as shown inFIG. 2.

FIG. 4 is a simplified diagram showing a power conversion systemaccording to an embodiment of the present invention.

FIG. 5 is a simplified diagram showing the controller as part of thepower conversion system as shown in FIG. 4 according to an embodiment ofthe present invention.

FIG. 6 is a simplified diagram showing the controller as part of thepower conversion system as shown in FIG. 4 according to anotherembodiment of the present invention.

FIG. 7 is a simplified diagram showing certain components of thecontroller as part of the power conversion system as shown in FIG. 4according to an embodiment of the present invention.

FIG. 8 is a simplified diagram showing certain components of thecontroller as part of the power conversion system as shown in FIG. 4according to another embodiment of the present invention.

FIG. 9 and FIG. 10 are simplified diagrams showing different operationconditions of the operational amplifier as part of the error amplifieras shown in FIG. 8 according to certain embodiments of the presentinvention.

FIG. 11 is a simplified diagram showing certain components of thecontroller as part of the power conversion system as shown in FIG. 4according to yet another embodiment of the present invention.

FIG. 12 is a simplified timing diagram for the system as shown in FIG. 1showing the difference between the actual peak magnitude of the primarycurrent and the desired peak magnitude as a function of the input linevoltage.

FIG. 13 is a simplified diagram showing certain components of the systemas shown in FIG. 4 according to another embodiment of the presentinvention.

FIG. 14 is a simplified timing diagram for the system as shown in FIG. 1showing the peak magnitude of the secondary current as a function of theoutput voltage.

FIG. 15 is a simplified diagram showing certain components of the systemas shown in FIG. 4 according to yet another embodiment of the presentinvention.

FIG. 16 is a simplified diagram showing certain components of the systemas shown in FIG. 4 according to yet another embodiment of the presentinvention.

FIG. 17 is a simplified diagram showing a controller as part of thepower conversion system as shown in FIG. 4 according to yet anotherembodiment of the present invention.

FIG. 18 is a simplified diagram showing a controller as part of thepower conversion system as shown in FIG. 4 according to yet anotherembodiment of the present invention.

FIG. 19 is a simplified diagram showing certain components of the systemas shown in FIG. 4 according to yet another embodiment of the presentinvention.

5. DETAILED DESCRIPTION OF THE INVENTION

The present invention is directed to integrated circuits. Moreparticularly, the invention provides a system and method for currentregulation. Merely by way of example, the invention has been applied topower conversion systems. But it would be recognized that the inventionhas a much broader range of applicability.

According to Equation 1, N, R_(cs) and V_(ref) _(_) _(ea) may affect theprecision of the constant output current. N and R_(cs) are parameters ofperipheral components, and can be selected through system design. Thus,the reference signal 222 affects significantly the precision of theconstant output current. In actual applications, the average outputcurrent and the turns ratio of the transformer 110 are oftenpredetermined, and thus the ratio between R_(cs) and V_(ref) _(_) _(ea)is also approximately fixed. Therefore, the larger the reference signal222 is in magnitude, the larger the resistance of the resistor 130. Alarge resistance of the resistor 130 often results in large power loss,and thus the reference signal 222 may need to have a small magnitude forbetter efficiency. But non-ideal factors (e.g., offset errors, gainerrors) may have larger negative effects on the precision of theconstant output current if the reference signal 222 has a smallmagnitude.

FIG. 4 is a simplified diagram showing a power conversion systemaccording to an embodiment of the present invention. This diagram ismerely an example, which should not unduly limit the scope of theclaims. One of ordinary skill in the art would recognize manyvariations, alternatives, and modifications. The system 400 includes acontroller 402, resistors 404, 424, 426 and 432, capacitors 406, 420 and434, a diode 408, a transformer 410 including a primary winding 412, asecondary winding 414 and an auxiliary winding 416, a power switch 428,a current sensing resistor 430, and a rectifying diode 418. Thecontroller 402 includes terminals 438, 440, 442, 444, 446 and 448. Forexample, the power switch 428 is a bipolar junction transistor. Inanother example, the power switch 428 is a MOS transistor. The system400 provides power to an output load 422, e.g., one or more LEDs.

According to one embodiment, an alternate-current (AC) input voltage 452is applied to the system 400. For example, a bulk voltage 450 (e.g., arectified voltage no smaller than 0 V) associated with the AC inputvoltage 452 is received by the resistor 404. In another example, thecapacitor 406 is charged in response to the bulk voltage 450, and avoltage 454 is provided to the controller 402 at the terminal 438 (e.g.,terminal VCC). In yet another example, if the voltage 454 is larger thana predetermined threshold voltage (e.g., a under-voltage lock-outthreshold) in magnitude, the controller 402 begins to operate normally,and outputs a driving signal 456 through the terminal 442 (e.g.,terminal GATE). In yet another example, the switch 428 is closed (e.g.,being turned on) or open (e.g., being turned off) in response to thedriving signal 456 so that the output current 458 is regulated to beapproximately constant.

According to another embodiment, the auxiliary winding 416 charges thecapacitor 406 through the diode 408 when the switch 428 is closed (e.g.,being turned on) in response to the driving signal 456 so that thecontroller 402 can operate normally. For example, a feedback signal 460is provided to the controller 402 through the terminal 440 (e.g.,terminal FB) in order to detect the ending of a demagnetization processof the secondary winding 418 for charging or discharging the capacitor434 using an internal error amplifier in the controller 402. In anotherexample, the resistor 430 is used for detecting a primary current 462flowing through the primary winding 412, and a current-sensing signal464 is provided to the controller 402 through the terminal 444 (e.g.,terminal CS) to be processed during each switching cycle. In yet anotherexample, peak magnitudes of the current-sensing signal 464 are sampledand provided to the internal error amplifier. In yet another example,the capacitor 420 is used to keep an output voltage 468 stable.

According to yet another embodiment, the controller 402 includes anerror amplifier (e.g., as shown in FIGS. 5-8 and 11) which, comparedwith the error amplifier 216, has higher output impedance and a largergain. For example, the controller 402 includes at least part of aline-voltage-compensation component (e.g., as shown in FIG. 13) forkeeping the output current approximately constant (e.g., with a smallerror) under a wide range of input line voltages. In another example,the resistor 432 is included in the line-voltage-compensation component.In yet another example, the controller 402 includes at least part of aload-compensation component (e.g., as shown in FIG. 15) for keeping theoutput current approximately constant (e.g., with a small error) under awide range of output voltages. In yet another example, the resistor 432is included in the load-compensation component. In some embodiments, theresistor 432 is omitted.

FIG. 5 is a simplified diagram showing the controller 402 as part of thepower conversion system 400 according to an embodiment of the presentinvention. This diagram is merely an example, which should not undulylimit the scope of the claims. One of ordinary skill in the art wouldrecognize many variations, alternatives, and modifications. Thecontroller 402 includes an oscillator 502, an under-voltage lock-out(UVLO) component 504, a modulation component 506, a logic controller508, a driving component 510, a demagnetization detector 512, an erroramplifier 516, a current-sensing component 514, and a chopping component540. In addition, the controller 402 includes at least part of aline-voltage-compensation component 550 and at least part of aload-compensation component 560.

According to one embodiment, the UVLO component 504 detects the signal454 and outputs a signal 518. For example, if the signal 454 is largerthan a predetermined threshold in magnitude, the controller 402 beginsto operate normally. In another example, if the signal 454 is smallerthan the predetermined threshold in magnitude, the controller 402 isturned off. In yet another example, the error amplifier 516 receives asignal 520 from the current-sensing component 514 and a reference signal522 and outputs an amplified signal 524 to the modulation component 506.In yet another example, the modulation component 506 also receives asignal 528 from the oscillator 502 and outputs a modulation signal 526.In yet another example, the signal 528 is a ramping signal andincreases, linearly or non-linearly, to a peak magnitude during eachswitching period. In yet another example, the modulation signal 526 is apulse-width-modulation (PWM) signal with a fixed switching frequency andthe duty cycle of the signal 526 is determined based on a comparisonbetween the signal 524 and the signal 528. In yet another example, thelogic controller 508 processes the modulation signal 526 and outputs acontrol signal 530 to the driving component 510 which generates thesignal 456 to turn on or off the switch 428. In yet another example, thelogic controller 508 also outputs the control signal 530 to the currentsensing component 514.

According to another embodiment, the control signal 456 is at a logichigh level if the control signal 530 is at a logic high level. Forexample, the control signal 456 is at a logic low level if the controlsignal 530 is at a logic low level. In another example, thedemagnetization detector 512 detects the feedback signal 460 and outputsa demagnetization signal 532 for determining the beginning and the endof the demagnetization process of the secondary winding 414 during eachswitching period. In yet another example, the demagnetization signal 532is at a logic high level during the demagnetization period of eachswitching cycle, and at a logic low level during the rest of eachswitching cycle. In yet another example, the demagnetization signal 532changes from the logic low level to the logic high level if the controlsignal 456 changes from the logic high level to the logic low level. Inyet another example, the demagnetization detector 512 outputs a samplingsignal 538 to the load-compensation component 560.

According to yet another embodiment, the chopping component 540 receivesa clock signal 534 from the oscillator 502 and outputs a signal 536 tothe error amplifier 516. For example, the signal 536 is also a clocksignal that has a 50% duty cycle and a frequency which is 1/N of thefrequency of the clock signal 534. In another example, if the signal 536is at a logic high level, the error amplifier 516 receives the referencesignal 522 at a non-inverting input terminal and receives the signal 520at an inverting input terminal; if the signal 536 is at a logic lowlevel, the error amplifier 516 receives the reference signal 522 at theinverting input terminal and receives the signal 520 at thenon-inverting input terminal. In response, the offset voltage of theerror amplifier 516 generated during the time period when the signal 536is at the logic high level, and the offset voltage of the erroramplifier 516 generated during the time period when the signal 536 is atthe logic low level are equal in magnitude but have differentpolarities, in certain embodiments. For example, if the signal 536 has a50% duty cycle, the offset voltage generated during the time period whenthe signal 536 is at the logic high level cancels out the offset voltagegenerated during the time period when the signal 536 is at the logic lowlevel. Thus, an average offset voltage during a complete chopping periodof the signal 536 is approximately zero, according to some embodiments.

In another embodiment, the line-voltage-compensation component 550 isused for keeping the output current approximately constant (e.g., with asmall error) under a wide range of input line voltages. For example, theload-compensation component 560 is used for keeping the output currentapproximately constant (e.g., with a small error) under a wide range ofoutput voltages.

As discussed above and further emphasized here, FIG. 5 is merely anexample, which should not unduly limit the scope of the claims. One ofordinary skill in the art would recognize many variations, alternatives,and modifications. For example, the current-sensing component 514 is thesame as the current-sensing component 214. In another example, the erroramplifier 516 can be replaced by the error amplifier 216. In someembodiments, the line-voltage-compensation component 550 is removed. Incertain embodiments, the load-compensation component 560 is removed.

FIG. 6 is a simplified diagram showing the controller 402 as part of thepower conversion system 400 according to another embodiment of thepresent invention. This diagram is merely an example, which should notunduly limit the scope of the claims. One of ordinary skill in the artwould recognize many variations, alternatives, and modifications. Thecontroller 402 includes a ramp-signal generator 602, an under-voltagelock-out (UVLO) component 604, a modulation component 606, a logiccontroller 608, a driving component 610, a demagnetization detector 612,an error amplifier 616, a current-sensing component 614, and a choppingcomponent 640. In addition, the controller 402 includes at least part ofa line-voltage-compensation component 650 and at least part of aload-compensation component 660.

According to one embodiment, the UVLO component 604 detects the signal454 and outputs a signal 618. For example, if the signal 454 is largerthan a predetermined threshold in magnitude, the controller 402 beginsto operate normally. In another example, if the signal 454 is smallerthan the predetermined threshold in magnitude, the controller 402 isturned off. In yet another example, the error amplifier 616 receives asignal 620 from the current-sensing component 614 and a reference signal622 and outputs an amplified signal 624 to the modulation component 606.In yet another example, the modulation component 606 also receives asignal 628 from the ramp-signal generator 602 and outputs a modulationsignal 626. In yet another example, the signal 628 is a ramping signaland increases, linearly or non-linearly, to a peak magnitude during eachswitching period. In yet another example, the modulation signal 626 doesnot have a fixed switching frequency. In yet another example, the logiccontroller 608 processes the modulation signal 626 and outputs a controlsignal 630 to the driving component 610 which generates the signal 456to turn on or off the switch 428. In yet another example, the logiccontroller 608 also outputs the control signal 630 to the currentsensing component 614.

According to another embodiment, the control signal 456 is at a logichigh level if the control signal 630 is at a logic high level. Forexample, the control signal 456 is at a logic low level if the controlsignal 630 is at a logic low level. In yet another example, thedemagnetization detector 612 detects the feedback signal 460 and outputsa demagnetization signal 632 for determining the beginning and the endof the demagnetization process of the secondary winding 414 during eachswitching period. In yet another example, the demagnetization signal 632is at a logic high level during the demagnetization period of eachswitching cycle, and at a logic low level during the rest of eachswitching cycle. In yet another example, the demagnetization signal 632changes from the logic low level to the logic high level if the controlsignal 456 changes from the logic high level to the logic low level. Inyet another example, the demagnetization detector 612 outputs a samplingsignal 638 to the load-compensation component 660. In yet anotherexample, the ramp-signal generator 602 receives the control signal 630and outputs the signal 628.

According to yet another embodiment, the chopping component 640 receivesthe control signal 630 from the logic controller 608 and outputs asignal 636 to the error amplifier 616. For example, the signal 636 isalso a clock signal that has a 50% duty cycle and a frequency which is1/N of the frequency of the control signal 630. In another example, thesignal 636 is used for chopping the error amplifier 616. In yet anotherexample, the line-voltage-compensation component 650 is used for keepingthe precision of the constant output current within a wide range ofinput line voltages. In yet another example, the load-compensationcomponent 660 is used for keeping the precision of the constant outputcurrent within a wide range of output voltages.

As discussed above and further emphasized here, FIG. 6 is merely anexample, which should not unduly limit the scope of the claims. One ofordinary skill in the art would recognize many variations, alternatives,and modifications. For example, the current-sensing component 614 is thesame as the current-sensing component 214. In another example, the erroramplifier 616 can be replaced by the error amplifier 216. In someembodiments, the line-voltage-compensation component 650 is removed. Incertain embodiments, the load-compensation component 660 is removed.

FIG. 7 is a simplified diagram showing certain components of thecontroller 402 as part of the power conversion system 400 according toan embodiment of the present invention. This diagram is merely anexample, which should not unduly limit the scope of the claims. One ofordinary skill in the art would recognize many variations, alternatives,and modifications. The controller 402 includes a current-sensingcomponent 714 and an error amplifier 716. The current-sensing component714 includes a switch 702 and a capacitor 704. The error amplifier 716includes operational amplifiers 710 and 712, a switch 706, transistors754, 756, 758 and 760, and resistors 750 and 752. For example, thecurrent-sensing component 714 is the same as the current-sensingcomponent 514, and the error amplifier 716 is the same as the erroramplifier 516, as shown in FIG. 5. In another example, thecurrent-sensing component 714 is the same as the current-sensingcomponent 614, and the error amplifier 716 is the same as the erroramplifier 616, as shown in FIG. 6.

According to one embodiment, the current-sensing component 714 samplesthe current-sensing signal 464 and outputs a signal 720, and the erroramplifier 716 amplifies the difference between a signal 720 and areference signal 722. For example, the switch 702 is closed (e.g., beingturned on) or open (e.g., being turned off) in response to a signal 788in order to sample peak magnitudes of the current-sensing signal 464 indifferent switching periods. In another example, the switch 702 isclosed if the signal 788 is at a logic high level, and the switch 702 isopened if the signal 788 is at a logic low level. In yet anotherexample, if the switch 702 is closed (e.g., being turned on) in responseto the signal 788, the capacitor 704 is charged and the signal 720increases in magnitude. In yet another example, during thedemagnetization process of the secondary winding 414, the signal 732 isat a logic high level, and the switch 706 is closed (e.g., being turnedon). In yet another example, a current 764 flowing through thetransistor 758 is proportional in magnitude to a current 766 flowingthrough the transistor 756.

According to another embodiment, the resistance of the resistor 750 islarger than the resistance of the resistor 752. For example, theresistance of the resistor 750 is equal to the resistance of theresistor 752 multiplied by a constant K (e.g., K>1). In another example,an average output current of the system 400 can be determined accordingto the following equation, without taking into account any errorcurrent:

$\begin{matrix}{\overset{\_}{I_{o}} = {\frac{1}{2} \times N \times \frac{V_{ref\_ ea}}{K \times R_{cs}}}} & \left( {{Equation}\mspace{14mu} 2} \right)\end{matrix}$where N represents a turns ratio between the primary winding 412 and thesecondary winding 414, V_(ref) _(_) _(ea) represents the referencesignal 722, and R_(cs) represents the resistance of the resistor 430.For example, without affecting the efficiency of the system 400 orchanging the resistance of the resistor 430, the reference signal 722can be increased in magnitude by increasing the resistance of theresistor 750, so that the negative effects caused by the offset voltageof the amplifier 710 may be reduced significantly. In another example,the signal 788, the signal 720, the signal 722, the signal 730, and thesignal 732 are the same as the signal 530, the signal 520, the signal522, the signal 530, and the signal 532, respectively. In yet anotherexample, the signal 788, the signal 720, the signal 722, the signal 730,and the signal 732 are the same as the signal 630, the signal 620, thesignal 622, the signal 630, and the signal 632, respectively.

FIG. 8 is a simplified diagram showing certain components of thecontroller 402 as part of the power conversion system 400 according toanother embodiment of the present invention. This diagram is merely anexample, which should not unduly limit the scope of the claims. One ofordinary skill in the art would recognize many variations, alternatives,and modifications. The controller 402 includes a current-sensingcomponent 814 and an error amplifier 816. The current-sensing component814 includes a switch 802 and a capacitor 804. The error amplifier 816includes operational amplifiers 810 and 812, a switch 806, transistors854, 856, 858 and 860, and resistors 850 and 852. For example, thecurrent-sensing component 814 is the same as the current-sensingcomponent 514, and the error amplifier 816 is the same as the erroramplifier 516, as shown in FIG. 5. In another example, thecurrent-sensing component 814 is the same as the current-sensingcomponent 614, and the error amplifier 816 is the same as the erroramplifier 616, as shown in FIG. 6.

According to one embodiment, the current-sensing component 814 samplesthe current-sensing signal 464 and outputs a signal 820, and the erroramplifier 816 amplifies the difference between a signal 820 and areference signal 822. For example, the switch 802 is closed (e.g., beingturned on) or open (e.g., being turned off) in response to a signal 888in order to sample peak magnitudes of the current-sensing signal 464 indifferent switching periods. In another example, the switch 802 isclosed if the signal 888 is at a logic high level, and the switch 802 isopened if the signal 888 is at a logic low level. In yet anotherexample, if the switch 802 is closed (e.g., being turned on) in responseto the signal 888, the capacitor 804 is charged and the signal 820increases in magnitude. In yet another example, during thedemagnetization process of the secondary winding 414, the signal 832 isat a logic high level, and the switch 806 is closed (e.g., being turnedon). In yet another example, a current 864 flowing through thetransistor 858 is proportional in magnitude to a current 866 flowingthrough the transistor 856.

According another embodiment, the resistance of the resistor 850 islarger than the resistance of the resistor 852. For example, theresistance of the resistor 850 is equal to the resistance of theresistor 852 multiplied by a constant K′ (e.g., K′>1). In anotherexample, an average output current of the system 400 can be determinedaccording to the following equation, without taking into account anyerror current:

$\begin{matrix}{\overset{\_}{I_{o}} = {\frac{1}{2} \times N \times \frac{V_{ref\_ ea}}{K^{\prime} \times R_{cs}}}} & \left( {{Equation}\mspace{14mu} 3} \right)\end{matrix}$where N represents a turns ratio between the primary winding 412 and thesecondary winding 414, V_(ref) _(_) _(ea) represents the referencesignal 822, and R_(cs) represents the resistance of the resistor 430.For example, without affecting the efficiency of the system 400 orchanging the resistance of the resistor 430, the reference signal 822can be increased in magnitude by increasing the resistance of theresistor 850, so that the negative effects caused by the offset of theamplifier 810 may be reduced significantly. For example, the signal 888,the signal 820, the signal 822, the signal 830, and the signal 832 arethe same as the signal 530, the signal 520, the signal 522, the signal530, and the signal 532, respectively. In yet another example, thesignal 888, the signal 820, the signal 822, the signal 830, and thesignal 832 are the same as the signal 630, the signal 620, the signal622, the signal 630, and the signal 632, respectively.

According yet another embodiment, the amplifier 812 receives a signal836 from a chopping component (e.g., the component 540, or the component640). For example, the status of the amplifier 812 changes in aswitching period in response to the signal 836 so that the negativeeffects of the offset voltage of the amplifier 812 can be reduced. Inanother example, the signal 836 is a logic signal with a 50% duty cycle,and is either at a logic high level (e.g., “1”) or at a logic low level(e.g., “0”). In another example, if the signal 836 is at a particularlogic level (e.g., “1”), the operational amplifier 812 has acorresponding status, and if the signal 836 is at another logic level(e.g., “0”), the operational amplifier 812 assumes a different status,as shown in FIG. 9 and FIG. 10.

FIG. 9 and FIG. 10 are simplified diagrams showing different operationconditions of the operational amplifier 812 as part of the erroramplifier 816 according to certain embodiments of the present invention.These diagrams are merely examples, which should not unduly limit thescope of the claims. One of ordinary skill in the art would recognizemany variations, alternatives, and modifications. The amplifier 812includes transistors 902, 904, 906, 916, 918, 924, 926, 928, 930, 936and 938, and switches 908, 910, 912, 914, 920, 922, 932 and 934. Inaddition, the amplifier 812 includes an inverting input terminal 940, anon-inverting input terminal 942 and an output terminal 952. Forexample, each of the switches 908, 910, 912, 914, 920, 922, 932 and 934can toggle between two states.

As shown in FIG. 9, in response to the signal 836 being at a particularlogic level (e.g., “1”), the operational amplifier 812 assumes a certainstatus, according to one embodiment. For example, a gate terminal of thetransistor 904 is connected to the inverting input terminal 940, and agate terminal of the transistor 906 is connected to the non-invertinginput terminal 942, in response to the states of the switches 908, 910,912 and 914. In another example, the transistor 918 and the transistor938 are in a same current path including the transistors 926 and 930 andthe output terminal 952, in response to the states of the switches 920,922, 932 and 934.

As shown in FIG. 10, in response to the signal 836 being at a differentlogic level (e.g., “0”), the operational amplifier 812 assumes adifferent status, according to another embodiment. For example, a gateterminal of the transistor 906 is connected to the inverting inputterminal 940, and a gate terminal of the transistor 904 is connected tothe non-inverting input terminal 942, in response to the states of theswitches 908, 910, 912 and 914. In another example, the transistor 916and the transistor 936 are in a same current path including thetransistors 926 and 930 and the output terminal 952, in response to thestates of the switches 920, 922, 932 and 934. As shown in FIG. 9 andFIG. 10, the transistor 902 provides a bias current 954 for the inputterminals 940 and 942, according to certain embodiments. For example,signals 944, 946, 948 and 950 are bias voltage signals.

As discussed above and further emphasized here, FIGS. 7 and 8 are merelyexamples, which should not unduly limit the scope of the claims. One ofordinary skill in the art would recognize many variations, alternatives,and modifications. For example, additional components/devices may beincluded in the error amplifier 716 or the error amplifier 816 to yieldlarge output impedance, to increase gain and/or to reduce the mismatchbetween the transistors 756 and 758 or the mismatch between thetransistors 856 and 858, respectively, as shown in FIG. 11.

FIG. 11 is a simplified diagram showing certain components of thecontroller 402 as part of the power conversion system 400 according toyet another embodiment of the present invention. This diagram is merelyan example, which should not unduly limit the scope of the claims. Oneof ordinary skill in the art would recognize many variations,alternatives, and modifications. The controller 402 includes acurrent-sensing component 1114 and an error amplifier 1116. Thecurrent-sensing component 1114 includes a switch 1102 and a capacitor1104. The error amplifier 1116 includes operational amplifiers 1110,1112 and 1170, a switch 1106, transistors 1154, 1156, 1158, 1160 and1172, and resistors 1150 and 1152. For example, the current-sensingcomponent 1114 is the same as the current-sensing component 514, and theerror amplifier 1116 is the same as the error amplifier 516. In anotherexample, the current-sensing component 1114 is the same as thecurrent-sensing component 614, and the error amplifier 1116 is the sameas the error amplifier 616.

According to one embodiment, the current-sensing component 1114 samplesthe current-sensing signal 464 and outputs a signal 1120, and the erroramplifier 1116 amplifies the difference between a signal 1120 and areference signal 1122. For example, the switch 1102 is closed (e.g.,being turned on) or open (e.g., being turned off) in response to asignal 1188 in order to sample peak magnitudes of the current-sensingsignal 464 in different switching periods. In another example, theswitch 1102 is closed if the signal 1188 is at a logic high level, andthe switch 1102 is opened if the signal 1188 is at a logic low level. Inyet another example, if the switch 1102 is closed (e.g., being turnedon) in response to the signal 1188, the capacitor 1104 is charged andthe signal 1120 increases in magnitude. In yet another example, duringthe demagnetization process of the secondary winding 414, the signal1132 is at a logic high level, and the switch 1106 is closed (e.g.,being turned on). In yet another example, the amplifier 1170 and thetransistor 1172 form a gain boost circuit. In yet another example, thegain boost circuit including the amplifier 1170 and the transistor 1172increases the output impedance of the error amplifier 1116, and makes avoltage 1166 at a drain terminal of the transistor 1156 approximatelyequal to a voltage 1168 at a drain terminal of the transistor 1158 inorder to reduce the mismatch between the transistor 1156 and 1158.

According to another embodiment, the resistance of the resistor 1150 islarger than the resistance of the resistor 1152. For example, theresistance of the resistor 1150 is equal to the resistance of theresistor 1152 multiplied by a constant K″ (e.g., K″>1). In anotherexample, an average output current of the system 400 can be determinedaccording to the following equation, without taking into account anyerror current:

$\begin{matrix}{\overset{\_}{I_{o}} = {\frac{1}{2} \times N \times \frac{V_{ref\_ ea}}{K^{''} \times R_{cs}}}} & \left( {{Equation}\mspace{14mu} 4} \right)\end{matrix}$where N represents a turns ratio between the primary winding 412 and thesecondary winding 414, V_(ref) _(_) _(ea) represents the referencesignal 1122, and R_(cs) represents the resistance of the resistor 430.For example, without affecting the efficiency of the system 400 orchanging the resistance of the resistor 430, the reference signal 1122can be increased in magnitude by increasing the resistance of theresistor 1150, so that the negative effects caused by the offset of theamplifier 1110 may be reduced significantly. For example, the signal1188, the signal 1120, the signal 1122, the signal 1130, and the signal1132 are the same as the signal 530, the signal 520, the signal 522, thesignal 530, and the signal 532, respectively. In yet another example,the signal 1188, the signal 1120, the signal 1122, the signal 1130, andthe signal 1132 are the same as the signal 630, the signal 620, thesignal 622, the signal 630, and the signal 632, respectively.

According yet another embodiment, the amplifier 1112 receives a signal1136 from a chopping component (e.g., the component 540, or thecomponent 640). For example, the status of the amplifier 1112 changes ina switching period in response to the signal 1136 so that the negativeeffects of the offset voltage of the amplifier 1112 can be reduced. Inanother example, the signal 1136 is a logic signal with a 50% dutycycle, and is either at a logic high level (e.g., “1”) or at a logic lowlevel (e.g., “0”). In another example, if the signal 1136 is at aparticular logic level (e.g., “1”), the operational amplifier 1112 has acorresponding status, and if the signal 1136 is at another logic level(e.g., “0”), the operational amplifier 1112 assumes a different status,similar to what are shown in FIG. 9 and FIG. 10.

Referring back to FIG. 1 and FIG. 2, the input line voltage 152 oftenvaries in a range 90 V˜264 V in actual applications. Due to non-idealfactors, such as transmission delay, the actual output current 158 isdifferent from the designed ideal current value, e.g., by an amountΔI_(out).

FIG. 12 is a simplified timing diagram for the system 100 showing thedifference between the actual peak magnitude of the primary current 162and the desired peak magnitude as a function of the input line voltage152. The waveform 1202 represents the signal 226 as a function of time,the waveform 1204 represents the signal 156 as a function of time, thewaveform 1206 represents the primary current 162 under a high input linevoltage as a function of time, and the waveform 1208 represents theprimary current 162 under a low input line voltage as a function oftime.

As shown in FIG. 12, during a switching period, the modulation component206 changes, at t₀, the signal 226 from a logic low level to a logichigh level (e.g., as shown by the waveform 1202 ), and the drivingcomponent 210 changes the signal 156 from the logic low level to thelogic high level (e.g., as shown by the waveform 1204). Then, at t₁, themodulation component 206 changes the signal 226 from the logic highlevel to the logic low level (e.g., as shown by the waveform 1202).After a transmission delay (e.g., T_(d)), the driving component 210changes the signal 156 from the logic high level to the logic low level(e.g., at t₂ as shown by the waveform 1204). Between t₀ and t₂, theprimary current 162 increases in magnitude, e.g., as shown by thewaveforms 1206 and 1208. The primary current associated with the highline input voltage increases faster than that associated with the lowline input voltage, e.g., as shown by the waveforms 1206 and 1208. Theincrease of the primary current 162 between t₁ and t₂ can be determinedaccording to the following equation:

$\begin{matrix}{{\Delta\; I} = {\frac{V_{AC}}{L} \times T_{d}}} & \left( {{Equation}\mspace{14mu} 5} \right)\end{matrix}$where V_(AC) represents the input line voltage 152, L represents theinductance of the primary winding 112, and T_(d) represents thetransmission delay. The increase of the current-sensing signal 164between t₁ and t₂ corresponding to the increase of the primary current162 is determined according to the following equation:

$\begin{matrix}{{\Delta\; V_{CS}} = {R\; 5 \times \frac{V_{AC}}{L} \times T_{d}}} & \left( {{Equation}\mspace{14mu} 6} \right)\end{matrix}$where R5 represents the resistance of the current sensing resistor 130.

According to Equation 5, with the same transmission delay, the change ofthe primary current 162 between t₁ and t₂ associated with the high lineinput voltage is larger in magnitude than that associated with the lowline input voltage. Thus, the residual amount of the output currentchanges with the line input voltage.

FIG. 13 is a simplified diagram showing certain components of the system400 according to another embodiment of the present invention. Thisdiagram is merely an example, which should not unduly limit the scope ofthe claims. One of ordinary skill in the art would recognize manyvariations, alternatives, and modifications. The controller 402 furtherincludes transistors 1304, 1306 and 1308, and an operational amplifier1310. For example, the line-voltage-compensation component 550 includesthe resistors 424, 430 and 432, the transistors 1304, 1306 and 1308, andthe operational amplifier 1310. In another example, theline-voltage-compensation component 650 includes the resistors 424, 430and 432, the transistors 1304, 1306 and 1308, and the operationalamplifier 1310.

According to one embodiment, a non-inverting input terminal (e.g., the“+” terminal as shown in FIG. 13) of the amplifier 1310 is connected toa ground voltage, and an inverting input terminal (e.g., the “−”terminal as shown in FIG. 13) of the amplifier 1310 is connected to theterminal 440 (e.g., terminal FB). For example, before and/or after thedemagnetization process, if the switch 428 is closed (e.g., being turnedon), a voltage 1318 at the auxiliary winding 416 is lower than theground voltage (e.g., 0 V). In another example, the operationalamplifier 1310 operates with the transistor 1308 to adjust a voltage1398 at the terminal 440 (e.g., FB) to be approximately equal to theground voltage (e.g., 0 V). In yet another example, a current 1316 flowsout of the controller 402 via the terminal 440 (e.g., FB) through theresistor 424. In another example, the current 1316 is determinedaccording to the following equation:

$\begin{matrix}{\;{I_{FB} = {\frac{V_{AC}}{R\; 6} \times N_{ap}}}} & \left( {{Equation}\mspace{14mu} 7} \right)\end{matrix}$where I_(FB) represents the current 1316, V_(AC) represents the lineinput voltage 452, N_(ap) represents the turns ratio between theauxiliary winding 416 and the primary winding 412, and R 6 representsthe resistance of the resistor 424. In yet another example, during thedemagnetization process, if the voltage 1318 at the auxiliary winding416 is higher than the ground voltage, a voltage 1398 at the terminal440 (e.g., FB) is higher than the ground voltage, and in response theoperational amplifier 1310 outputs a signal 1396 to turn off thetransistor 1308.

According to another embodiment, the size (e.g., the ratio between thewidth and the length) of the transistor 1304 is proportional to that ofthe transistor 1306. For example, a current mirror circuit including thetransistors 1304 and 1306 mirrors the current 1316 to generate a current1314. In another example, the current 1314 flows from the transistor1306 to the resistor 432 through the terminal 444 (e.g., terminal CS).In yet another example, the current 1314 before and/or after thedemagnetization process is determined according to the followingequation:

$\begin{matrix}{\;{I_{CC\_ AC} = {\frac{V_{AC}}{R\; 6} \times N_{ap} \times \frac{1}{s}}}} & \left( {{Equation}\mspace{14mu} 8} \right)\end{matrix}$where I_(CC) _(_) _(AC) represents the current 1314, and s representsthe ratio between the size of the transistor 1304 and that of thetransistor 1306. In another example, a compensation value for acurrent-sensing signal 1320 before and/or after the demagnetizationprocess is determined according to the following equation:

$\begin{matrix}{{\Delta\;{CS}} = {\frac{R\; 4 \times V_{A\; C}}{R\; 6} \times N_{ap} \times \frac{1}{s}}} & \left( {{Equation}\mspace{14mu} 9} \right)\end{matrix}$where ΔCS represents the compensation value for the current-sensingsignal 1320 and R4 represents the resistance of the resistor 432. Forexample, the voltage at the terminal 444 is equal to the current-sensingsignal 1320 raised by ΔCS in magnitude. In another example, thecurrent-sensing signal 1320 is proportional to the primary current 462in magnitude.

In one embodiment, if certain error output current is taken into accountbut no compensation for the current-sensing signal 1320 (e.g., ΔCS asshown in Equation 9) is taken into account, an average output current ofthe system 400 is determined according to the following equation:

$\begin{matrix}{\overset{\_}{I_{o}} = {{\frac{1}{2} \times N \times \frac{V_{ref\_ ea}}{R_{cs}}} + {\Delta\;{I_{o}\left( V_{A\; C} \right)}}}} & \left( {{Equation}\mspace{14mu} 10} \right)\end{matrix}$where N represents a turns ratio between the primary winding 412 and thesecondary winding 414, V_(ref) _(_) _(ea) represents an internalreference signal (e.g., the reference signal 522, or the referencesignal 622 ), R_(cs) represents the resistance of the resistor 430, andΔI_(o)(V_(AC)) represents an error output current as a function of theline input voltage 452 (e.g., V_(AC)).

In another embodiment, if a compensation for the current-sensing signal1320 (e.g., ΔCS as shown in Equation 9) is also taken into account, theinternal reference signal (e.g., the reference signal 522, or thereference signal 622) is effectively reduced by the same amount ofcompensation. For example, based on Equation 10, the following isobtained:

$\begin{matrix}{\overset{\_}{I_{o}} = {{\frac{1}{2} \times N \times \frac{V_{ref\_ ea} - {\Delta\;{CS}}}{R_{cs}}} + {\Delta\;{I_{o}\left( V_{A\; C} \right)}}}} & \left( {{Equation}\mspace{14mu} 11} \right)\end{matrix}$Thus, under different input line voltages, the average output current ofthe system 400 can be kept approximately constant by adjusting thecompensation value ΔCS for the current-sensing signal 1320 (e.g., byadjusting the resistances of the resistors 432 and 424) according tocertain embodiments.

Referring back to FIG. 1 and FIG. 2, oftentimes, because of the leakageinductance of the transformer 110, the energy in the primary side cannotbe fully transferred to the secondary side, and thus the output current158 is different from the designed ideal current value, e.g., by anamount ΔI_(out). Such difference changes with the output voltage 168.

FIG. 14 is a simplified timing diagram for the system 100 showing thepeak magnitude of the secondary current 169 as a function of the outputvoltage 168. The waveform 1402 represents the primary current 162 as afunction of time, and the waveform 1404 represents the secondary current169 as a function of time. For example, the secondary current 169 isclosely related to the output current 158.

As shown in FIG. 14, during an on-time period (e.g., between t₅ and t₆),the switch 128 is closed (e.g., being turned on), and the primarycurrent 162 increases in magnitude, e.g., as shown by the waveform 1402.At t₆, the switch 128 is open (e.g., being turned off). During a timeperiod Δt (e.g., between t₆ and t₇), the primary current 162 decreasesin magnitude due to the leakage inductance (e.g., as shown by thewaveform 1402 ), and the secondary current 169 increases in magnitudefrom a low magnitude (e.g., t₆) to a peak magnitude (e.g., at t₇) asshown by the waveform 1404. After t₇, the secondary current 169decreases in magnitude, e.g., as shown by the waveform 1404. If theoutput voltage 168 increases in magnitude, the duration of the timeperiod Δt (e.g., between t₆ and t₇) increases and the peak magnitude ofthe secondary current 169 decreases (e.g., as shown by the waveform1404). Thus, the output current 158 decreases in magnitude, and thedifference between the output current 158 and the ideal current valueincreases.

FIG. 15 is a simplified diagram showing certain components of the system400 according to yet another embodiment of the present invention. Thisdiagram is merely an example, which should not unduly limit the scope ofthe claims. One of ordinary skill in the art would recognize manyvariations, alternatives, and modifications. The controller 402 furtherincludes transistors 1504, 1506, 1510, 1518 and 1520, an operationalamplifier 1508, a resistor 1516, a capacitor 1512, and a switch 1514.For example, the load-compensation component 560 includes the resistor432, the transistors 1504, 1506, 1510, 1518 and 1520, the operationalamplifier 1508, the resistor 1516, the capacitor 1512, and the switch1514. In another example, the load-compensation component 660 includesthe resistor 432, the transistors 1504, 1506, 1510, 1518 and 1520, theoperational amplifier 1508, the resistor 1516, the capacitor 1512, andthe switch 1514.

According to one embodiment, if the switch 1514 is closed (e.g., beingturned on) in response to a sampling signal 1538 during thedemagnetization process of the secondary winding 414, the feedbacksignal 460 is sampled and held at the capacitor 1512. For example, theamplifier 1508 receives the sampled-and-held signal 1530 at anon-inverting input terminal (e.g., the “+” terminal as shown in FIG.15). In another example, the sampled-and-held signal 1530 is determinedaccording to the following equation:V _(FB) _(_) _(sample) =V _(out) ×N _(as)  (Equation 13)where V_(FB) _(_) _(sample) represents the sampled-and-held signal 1530,V_(out) represents the output voltage 468, and N_(as) represents theturns ratio between the auxiliary winding 416 and the secondary winding414. In yet another example, before and/or after the demagnetizationprocess of the secondary winding 414, the switch 1514 is opened (e.g.,being turned off) in response to the sampling signal 1538. In yetanother example, the sampling signal 1538 is the same as the signal 538.In yet another example, the sampling signal 1538 is the same as thesignal 638.

According to another embodiment, if the switch 428 is closed (e.g.,being turned on) during an on-time period, a current 1522 flows throughthe transistors 1504 and 1510 and the resistor 1516. For example, acurrent-mirror circuit including the transistors 1504, 1506, 1518 and1520 mirrors the current 1522 to generate a current 1524 that flowsthrough the transistors 1506 and 1518, and mirrors the current 1524 togenerate a compensation current 1526 (e.g., I_(CC) _(_) _(load)). Inanother example, the compensation current 1526 flows from the resistor432 to the transistor 1520 through the terminal 444. In yet anotherexample, during the demagnetization process, the compensation current1526 is determined according to the following equation:

$\begin{matrix}{I_{CC\_ load} = {\frac{V_{FB\_ sample}}{R\; 151} \times \frac{1}{p}}} & \left( {{Equation}\mspace{14mu} 14} \right)\end{matrix}$where I_(CC) _(_) _(load) represents the current 1526, V_(FB) _(_)_(sample) represents the sampled-and-held signal 1530, R151 representsthe resistance of the resistor 1516, and p represents a ratio associatedwith the current mirror circuit including the transistors 1504, 1506,1518 and 1520. In yet another example, the current 1526 flows throughthe resistor 432 to provide a compensation value for a current-sensingsignal 1532 which, during the demagnetization process, can be determinedaccording to the following equation:

$\begin{matrix}{{\Delta\;{CS}^{\prime}} = {\frac{R\; 4 \times V_{FB\_ sample}}{R\; 151} \times \frac{1}{p}}} & \left( {{Equation}\mspace{14mu} 15} \right)\end{matrix}$where ΔCS′ represents the compensation value for the current-sensingsignal 1532 and R4 represents the resistance of the resistor 432. Forexample, the voltage at the terminal 444 is equal to the current-sensingsignal 1532 lowered by ΔCS′ in magnitude. In another example, thecurrent-sensing signal 1532 is proportional to the primary current 462in magnitude.

In one embodiment, if certain error output current is taken into accountbut no compensation for the current-sensing signal 1532 (e.g., ΔCS′ asshown in Equation 15) is taken into account, an average output currentof the system 400 is determined according to the following equation:

$\begin{matrix}{\overset{\_}{I_{o}} = {{\frac{1}{2} \times N \times \frac{V_{ref\_ ea}}{R_{cs}}} + {\Delta\;{I_{o}\left( V_{out} \right)}}}} & \left( {{Equation}\mspace{14mu} 16} \right)\end{matrix}$where N represents a turns ratio between the primary winding 412 and thesecondary winding 414, V_(ref) _(_) _(ea) represents an internalreference signal (e.g., the reference signal 522, or the referencesignal 622 ), R_(cs) represents the resistance of the resistor 430, andΔI_(o)(V_(out)) represents an error current as a function of the outputvoltage 468 (e.g., V_(out)).

In another embodiment, if a compensation for the current-sensing signal1532 (e.g., ΔCS′ as shown in Equation 15) is also taken into account,the internal reference signal (e.g., the reference signal 522, or thereference signal 622) is effectively reduced by the same amount ofcompensation. For example, based on Equation 16, the following isobtained:

$\begin{matrix}{\overset{\_}{I_{o}} = {{\frac{1}{2} \times N \times \frac{V_{ref\_ ea} - {\Delta\;{CS}^{\prime}}}{R_{cs}}} + {\Delta\;{I_{o}\left( V_{out} \right)}}}} & \left( {{Equation}\mspace{14mu} 17} \right)\end{matrix}$Thus, under different output voltages, the average output current of thesystem 400 can be kept approximately constant by adjusting thecompensation value ΔCS′ for the current-sensing signal 1532 (e.g., byadjusting the resistances of the resistors 432 and 1516) according tocertain embodiments.

FIG. 16 is a simplified diagram showing certain components of the system400 according to yet another embodiment of the present invention. Thisdiagram is merely an example, which should not unduly limit the scope ofthe claims. One of ordinary skill in the art would recognize manyvariations, alternatives, and modifications. The controller 402 furtherincludes transistors 1604, 1606, 1618 and 1620, a transconductanceamplifier 1608, a capacitor 1612, and a switch 1614. For example, theload-compensation component 560 includes the resistor 432, thetransistors 1604, 1606, 1618 and 1620, the transconductance amplifier1608, the capacitor 1612, and the switch 1614. in another example, theload-compensation component 660 includes the resistor 432, thetransistors 1604, 1606, 1618 and 1620, the transconductance amplifier1608, the capacitor 1612, and the switch 1614.

According to one embodiment, if the switch 1614 is closed (e.g., beingturned on) in response to a sampling signal 1638 during thedemagnetization process of the secondary winding 414, the feedbacksignal 460 is sampled and held at the capacitor 1612. For example, thetransconductance amplifier 1608 receives the sampled-and-held signal1630 at an inverting input terminal (e.g., the “−” terminal as shown inFIG. 16) and a threshold signal 1610 (e.g., V_(th) _(_) _(load)) at anon-inverting input terminal (e.g., the “+” terminal as shown in FIG.16) and generates a current 1622 (e.g., I_(gm)). In another example, thesampled-and-held signal 1630 is determined according to the followingequation:V _(FB) _(_) _(sample) =V _(out) ×N _(as)  (Equation 18)where V_(FB) _(_) _(sample) represents the sampled-and-held signal 1630,V_(out) represents the output voltage 468, and N_(as) represents theturns ratio between the auxiliary winding 416 and the secondary winding414. In yet another example, before and/or after the demagnetizationprocess of the secondary winding 414, the switch 1614 is opened (e.g.,being turned off) in response to the sampling signal 1638. In yetanother example, the sampling signal 1638 is the same as the signal 538.In yet another example, the sampling signal 1638 is the same as thesignal 638.

According to another embodiment, a current-mirror circuit including thetransistors 1504, 1506, 1518 and 1520 mirrors the current 1622 togenerate a current 1624 that flows through the transistors 1604 and1620, and mirrors the current 1624 to generate a compensation current1626 (e.g., I_(CC) _(_) _(load)). For example, the compensation current1626 flows from the transistor 1606 to the resistor 432 through theterminal 444. In another example, during the demagnetization process,the compensation current 1626 is determined according to the followingequation:

$\begin{matrix}{I_{CC\_ load} = {\left( {V_{th\_ load} - V_{FB\_ sample}} \right) \times g_{m} \times \frac{1}{p}}} & \left( {{Equation}\mspace{14mu} 19} \right)\end{matrix}$where I_(CC) _(_) _(load) represents the current 1626, V_(FB) _(_)_(sample) represents the sampled-and-held signal 1630, g_(m) representsthe transconductance of the amplifier 1608, and p represents a ratioassociated with the current mirror circuit including the transistors1604, 1606, 1618 and 1620. In another example, the compensation current1626 (e.g., I_(CC) _(_) _(load)) flows through the resistor 432 toprovide a compensation value for a current-sensing signal 1632 which,during the demagnetization process, can be determined according to thefollowing equation:

$\begin{matrix}{{\Delta\;{CS}^{\;''}} = {R\; 4 \times \left( {V_{th\_ load} - V_{FB\_ sample}} \right) \times g_{m} \times \frac{1}{p}}} & \left( {{Equation}\mspace{14mu} 20} \right)\end{matrix}$where ΔCS″ represents the compensation value for the current-sensingsignal 1632, and R4 represents the resistance of the resistor 432. Forexample, the voltage at the terminal 444 is equal to the current-sensingsignal 1632 raised by ΔCS″ in magnitude. In another example, thecurrent-sensing signal 1632 is proportional to the primary current 462in magnitude.

In one embodiment, if certain error output current is taken into accountbut no compensation for the current-sensing signal 1632 (e.g., ΔCS″ asshown in Equation 20) is taken into account, an average output currentof the system 400 is determined according to the following equation:

$\begin{matrix}{\overset{\_}{I_{o}} = {{\frac{1}{2} \times N \times \frac{V_{ref\_ ea}}{R_{cs}}} + {\Delta\;{I_{o}\left( V_{out} \right)}}}} & \left( {{Equation}\mspace{14mu} 21} \right)\end{matrix}$where N represents a turns ratio between the primary winding 412 and thesecondary winding 414, V_(ref) _(_) _(ea) represents an internalreference signal (e.g., the reference signal 522, or the referencesignal 622), R_(cs) represents the resistance of the resistor 430, andΔI_(o)(V_(out)) represents an error current as a function of the outputvoltage 468 (e.g., V_(out)).

In another embodiment, if a compensation for the current-sensing signal1632 (e.g., ΔCS″ as shown in Equation 20) is also taken into account,the internal reference signal (e.g., the reference signal 522, or thereference signal 622) is effectively reduced by the same amount ofcompensation. For example, based on Equation 21, the following isobtained:

$\begin{matrix}{\overset{\_}{I_{o}} = {{\frac{1}{2} \times N \times \frac{V_{ref\_ ea} - {\Delta\;{CS}^{\;''}}}{R_{cs}}} + {\Delta\;{I_{o}\left( V_{out} \right)}}}} & \left( {{Equation}\mspace{14mu} 22} \right)\end{matrix}$Thus, under different output voltages, the average output current of thesystem 400 can be kept approximately constant by adjusting thecompensation value ΔCS″ for the current-sensing signal 1632 (e.g.,adjusting the resistance of the resistor 432), according to certainembodiments.

FIG. 17 is a simplified diagram showing the controller 402 as part ofthe power conversion system 400 according to yet another embodiment ofthe present invention. This diagram is merely an example, which shouldnot unduly limit the scope of the claims. One of ordinary skill in theart would recognize many variations, alternatives, and modifications.

The controller 402 includes an oscillator 1702, an under-voltagelock-out (UVLO) component 1704, a modulation component 1706, a logiccontroller 1708, a driving component 1710, a demagnetization detector1712, an error amplifier 1716, a current-sensing component 1714, and achopping component 1740. In addition, the controller 402 includes atleast part of a line-voltage-compensation component 1750 and at leastpart of a load-compensation component 1760. For example, the erroramplifier 1716 is the same as the error amplifier 516. In anotherexample, the line-voltage-compensation component 1750 is the same as theline-voltage-compensation component 550. In yet another example, theload-compensation component 1760 is the same as the load-compensationcomponent 560.

According to one embodiment, the UVLO component 1704 detects the signal454 and outputs a signal 1718. For example, if the signal 454 is largerthan a predetermined threshold in magnitude, the controller 402 beginsto operate normally. In another example, if the signal 454 is smallerthan the predetermined threshold in magnitude, the controller 402 isturned off. In yet another example, the error amplifier 1716 receives asignal 1720 from the current-sensing component 1714 and a referencesignal 1722 and outputs an amplified signal 1724 to the modulationcomponent 1706. In yet another example, the modulation component 1706also receives a signal 1728 from the oscillator 1702 and outputs amodulation signal 1726. In yet another example, the signal 1728 is aramping signal and increases, linearly or non-linearly, to a peakmagnitude during each switching period. In yet another example, themodulation signal 1726 is a pulse-width-modulation (PWM) signal with afixed switching frequency and the duty cycle of the signal 1726 isdetermined based on a comparison between the signal 1724 and the signal1728. In yet another example, the logic controller 1708 processes themodulation signal 1726 and outputs a control signal 1730 to the drivingcomponent 1710 which generates the signal 456 to turn on or off theswitch 428. In yet another example, the logic controller 1708 alsooutputs the control signal 1730 to the current sensing component 1714.

According to another embodiment, the control signal 456 is at a logichigh level if the control signal 1730 is at a logic high level. Forexample, the control signal 456 is at a logic low level if the controlsignal 1730 is at a logic low level. In another example, thedemagnetization detector 1712 detects the feedback signal 460 andoutputs a demagnetization signal 1732 for determining the beginning andthe end of the demagnetization process of the secondary winding 414during each switching period. In yet another example, thedemagnetization signal 1732 is at a logic high level during thedemagnetization period of each switching cycle, and at a logic low levelduring the rest of each switching cycle. In yet another example, thedemagnetization signal 1732 changes from the logic low level to thelogic high level if the control signal 456 changes from the logic highlevel to the logic low level. In yet another example, thedemagnetization detector 1712 outputs a sampling signal 1738 to theload-compensation component 1760 which outputs a compensation signal1798 to affect the output of the error amplifier 1716.

According to yet another embodiment, the chopping component 1740receives a clock signal 1734 from the oscillator 1702 and outputs asignal 1736 to the error amplifier 1716. For example, the signal 1736 isalso a clock signal that has a 50% duty cycle and a frequency which is1/N of the frequency of the clock signal 1734. In another example, thesignal 1736 is used for chopping the error amplifier 1716. In yetanother example, the line-voltage-compensation component 1750 is usedfor keeping the output current approximately constant (e.g., with asmall error) under a wide range of input line voltages. In yet anotherexample, the load-compensation component 1760 is used for keeping theoutput current approximately constant (e.g., with a small error) under awide range of output voltages.

As discussed above and further emphasized here, FIG. 17 is merely anexample, which should not unduly limit the scope of the claims. One ofordinary skill in the art would recognize many variations, alternatives,and modifications. For example, the current-sensing component 1714 isthe same as the current-sensing component 214. In another example, theerror amplifier 1716 can be replaced by the error amplifier 216. In someembodiments, the line-voltage-compensation component 1750 is removed. Incertain embodiments, the load-compensation component 1760 is removed.

FIG. 18 is a simplified diagram showing the controller 402 as part ofthe power conversion system 400 according to yet another embodiment ofthe present invention. This diagram is merely an example, which shouldnot unduly limit the scope of the claims. One of ordinary skill in theart would recognize many variations, alternatives, and modifications.

The controller 402 includes a ramp-signal generator 2002, anunder-voltage lock-out (UVLO) component 2004, a modulation component2006, a logic controller 2008, a driving component 2010, ademagnetization detector 2012, an error amplifier 2016, acurrent-sensing component 2014, and a chopping component 2040. Inaddition, the controller 402 includes at least part of aline-voltage-compensation component 2050 and at least part of aload-compensation component 2060. For example, the error amplifier 2016is the same as the error amplifier 616. In another example, theline-voltage-compensation component 2050 is the same as theline-voltage-compensation component 650. In yet another example, theload-compensation component 2060 is the same as the load-compensationcomponent 660.

According to one embodiment, the UVLO component 2004 detects the signal454 and outputs a signal 2018. For example, if the signal 454 is largerthan a predetermined threshold in magnitude, the controller 402 beginsto operate normally. In another example, if the signal 454 is smallerthan the predetermined threshold in magnitude, the controller 402 isturned off. In yet another example, the error amplifier 2016 receives asignal 2020 from the current-sensing component 2014 and a referencesignal 2022 and outputs an amplified signal 2024 to the modulationcomponent 2006. In yet another example, the modulation component 2006also receives a signal 2028 from the ramp-signal generator 2002 andoutputs a modulation signal 2026. In yet another example, the signal2028 is a ramping signal and increases, linearly or non-linearly, to apeak magnitude during each switching period. In yet another example, themodulation signal 2026 does not have a fixed switching frequency. In yetanother example, the logic controller 2008 processes the modulationsignal 2026 and outputs a control signal 2030 to the driving component2010 which generates the signal 456 to turn on or off the switch 428. Inyet another example, the logic controller 2008 also outputs the controlsignal 2030 to the current sensing component 2014.

According to another embodiment, the control signal 456 is at a logichigh level if the control signal 2030 is at a logic high level. Forexample, the control signal 456 is at a logic low level if the controlsignal 2030 is at a logic low level. In another example, thedemagnetization detector 2012 detects the feedback signal 2060 andoutputs a demagnetization signal 2032 for determining the beginning andthe end of the demagnetization process of the secondary winding 414during each switching period. In yet another example, thedemagnetization signal 2032 is at a logic high level during thedemagnetization period of each switching cycle, and at a logic low levelduring the rest of each switching cycle. In yet another example, thedemagnetization signal 2032 changes from the logic low level to thelogic high level if the control signal 456 changes from the logic highlevel to the logic low level. In yet another example, thedemagnetization detector 2012 outputs a sampling signal 2038 to theload-compensation component 2060 which outputs a compensation signal2098 to affect the output of the error amplifier 2016. In yet anotherexample, the ramp-signal generator 2002 receives the control signal 2030and outputs the signal 2028 to the modulation component 2006.

According to yet another embodiment, the chopping component 2040receives the control signal 2030 from the logic controller 2008 andoutputs a signal 2036 to the error amplifier 2016. For example, thesignal 2036 is a clock signal that has a 50% duty cycle and a frequencywhich is 1/N of the frequency of the control signal 2030. In anotherexample, the signal 2036 is used for chopping the error amplifier 2016.In yet another example, the line-voltage-compensation component 2050 isused for keeping the precision of the constant output current within awide range of input line voltages. In yet another example, theload-compensation component 2060 is used for keeping the precision ofthe constant output current within a wide range of output voltages.

As discussed above and further emphasized here, FIG. 18 is merely anexample, which should not unduly limit the scope of the claims. One ofordinary skill in the art would recognize many variations, alternatives,and modifications. For example, the current-sensing component 2014 isthe same as the current-sensing component 214. In another example, theerror amplifier 2016 can be replaced by the error amplifier 216. In someembodiments, the line-voltage-compensation component 2050 is removed. Incertain embodiments, the load-compensation component 2060 is removed.

FIG. 19 is a simplified diagram showing certain components of the system400 according to yet another embodiment of the present invention. Thisdiagram is merely an example, which should not unduly limit the scope ofthe claims. One of ordinary skill in the art would recognize manyvariations, alternatives, and modifications.

The controller 402 includes a load-compensation component 1960, acurrent-sensing component 1914, and an error amplifier 1916. Theload-compensation component 1960 includes transistors 1818 and 1820, atransconductance amplifier 1808, a capacitor 1812, and a switch 1814.For example, the load-compensation component 1960 is the same as theload-compensation component 1760. In another example, theload-compensation component 1960 is the same as the load-compensationcomponent 2060. In yet another example, the current-sensing component1914 is the same as the current-sensing component 1714. In yet anotherexample, the current-sensing component 1914 is the same as thecurrent-sensing component 2014. In yet another example, the erroramplifier 1916 is the same as the error amplifier 1716. In yet anotherexample, the error amplifier 1916 is the same as the error amplifier2016.

According to one embodiment, an output current 1924 of the erroramplifier 1916 is related to a difference between a reference signal1922 and an output signal 1920 generated by the current sensingcomponent 1914. For example, the output current 1924 of the erroramplifier 1916 is proportional to a difference between the referencesignal 1922 and the output signal 1920 in magnitude.

According to another embodiment, if the switch 1814 is closed (e.g.,being turned on) in response to the sampling signal 1938 during thedemagnetization process of the secondary winding 414, the feedbacksignal 460 is sampled and held at the capacitor 1812. For example, thetransconductance amplifier 1808 receives the sampled-and-held signal1830 at an inverting input terminal (e.g., the “−” terminal as shown inFIG. 19) and a threshold signal 1810 (e.g., V_(th) _(_) _(load)) at anon-inverting input terminal (e.g., the “+” terminal as shown in FIG.19) and generates a current 1822 (e.g., I_(gm)). In another example, thesampled-and-held signal 1830 is determined according to the followingequation:V _(FB) _(_) _(sample) =V _(out) ×N _(as)  (Equation 23)where V_(FB) _(_) _(sample) represents the sampled-and-held signal 1830,V_(out) represents the output voltage 468, and N_(as) represents theturns ratio between the auxiliary winding 416 and the secondary winding414. In yet another example, before and/or after the demagnetizationprocess of the secondary winding 414, the switch 1814 is opened (e.g.,being turned off) in response to the sampling signal 1938. In yetanother example, the sampling signal 1938 is the same as the signal1738. In yet another example, the sampling signal 1938 is the same asthe signal 2038.

According to yet another embodiment, a current-mirror circuit includingthe transistors 1818 and 1820 mirrors the current 1822 to generate acompensation current 1824 (e.g., I_(CC) _(_) _(load)). For example, thecurrent-mirror circuit is coupled to the error amplifier 1916 at acircuit node 1930, and the output current 1924 is divided into thecompensation current 1824 and a current 1926. In one embodiment, theoutput current 1924 flows from the error amplifier 1916 to the circuitnode 1930, the compensation current flows from the circuit node 1930 tothe transistor 1820, and the current 1926 flows from the circuit node1930 to the terminal 448. In another embodiment, the current 1926 isequal to the output current 1924 minus the compensation current 1824 inmagnitude.

According to yet another embodiment, during the demagnetization process,the compensation current 1824 is determined according to the followingequation:

$\begin{matrix}{I_{CC\_ load} = {\left( {V_{th\_ load} - V_{FB\_ sample}} \right) \times g_{m} \times \frac{1}{p}}} & \left( {{Equation}\mspace{14mu} 24} \right)\end{matrix}$where I_(CC) _(_) _(load) represents the current 1824, V_(FB) _(_)_(sample) represents the sampled-and-held signal 1830, g_(m) representsthe transconductance of the amplifier 1808, and p represents a ratioassociated with the current minor circuit including the transistors 1818and 1820. For example, the effect of the compensation current 1824(e.g., I_(CC) _(_) _(load)) on the error amplifier 1916, during thedemagnetization process, is approximately equivalent to generating acompensation value of a reference signal 1922 (e.g., V_(ref) _(_)_(ea)). In another example, the compensation value of the referencesignal 1922 is determined according to the following equation:

$\begin{matrix}{{\Delta\; V_{ref\_ ea}} = {\frac{1}{g_{m\_ ea}} \times \left( {V_{th\_ load} - V_{FB\_ sample}} \right) \times g_{m} \times \frac{1}{p}}} & \left( {{Equation}\mspace{14mu} 25} \right)\end{matrix}$where ΔV_(ref) _(_) _(ea) represents the compensation value for thereference signal 1922, and g_(m) _(_) _(ea) represents atransconductance of the error amplifier 1916.

In one embodiment, if certain error output current is taken into accountbut no compensation for the reference signal 1922 (e.g., V_(ref) _(_)_(ea)) (e.g., ΔV_(ref) _(_) _(ea) as shown in Equation 25) is taken intoaccount, an average output current of the system 400 is determinedaccording to the following equation:

$\begin{matrix}{\overset{\_}{I_{o}} = {{\frac{1}{2} \times N \times \frac{V_{ref\_ ea}}{R_{cs}}} + {\Delta\;{I_{o}\left( V_{out} \right)}}}} & \left( {{Equation}\mspace{14mu} 26} \right)\end{matrix}$where N represents a turns ratio between the primary winding 412 and thesecondary winding 414, V_(ref) _(_) _(ea) represents the referencesignal 1922, R_(cs) represents the resistance of the resistor 430, andΔI_(o)(V_(out)) represents an error current as a function of the outputvoltage 468 (e.g., V_(out).

In another embodiment, if a compensation for the reference signal 1922(e.g., V_(ref) _(_) _(ea)) (e.g., ΔV_(ref) _(_) _(ea) as shown inEquation 25) is also taken into account, an average output current ofthe system 400 is determined according to the following equation:

$\begin{matrix}\begin{matrix}{\overset{\_}{I_{o}} = {{\frac{1}{2} \times N \times \frac{V_{ref\_ ea} + {\Delta\; V_{ref\_ ea}}}{R_{cs}}} + {\Delta\;{I_{o}\left( V_{out} \right)}}}} & \;\end{matrix} & \left( {{Equation}\mspace{14mu} 27} \right)\end{matrix}$

Thus, under different output voltages, the average output current of thesystem 400 can be kept approximately constant by adjusting thecompensation value (e.g., ΔV_(ref) _(_) _(ea)) for the reference signal1922 (e.g., adjusting the ratio between the resistance of the resistor424 and the resistance of the resistor 426), according to certainembodiments.

According to another embodiment, an error amplifier for processing areference signal and an input signal associated with a current of apower conversion system includes a first operational amplifier, a secondoperational amplifier, a first transistor, a second transistor, acurrent minor component, a switch, a first resistor and a secondresistor. The first operational amplifier includes a first inputterminal, a second input terminal and a first output terminal, the firstinput terminal being configured to receive a reference signal. The firsttransistor includes a first transistor terminal, a second transistorterminal and a third transistor terminal, the first transistor terminalbeing configured to receive a first amplified signal from the firstoutput terminal, the third transistor terminal being coupled to thesecond input terminal. The second operational amplifier includes a thirdinput terminal, a fourth input terminal and a second output terminal,the third input terminal being configured to receive an input signalassociated with a current flowing through a primary winding of a powerconversion system. The second transistor includes a fourth transistorterminal, a fifth transistor terminal and a sixth transistor terminal,the fourth transistor terminal being configured to receive a secondamplified signal from the second output terminal, the sixth transistorterminal being coupled to the fourth input terminal. The current mirrorcomponent includes a first component terminal and a second componentterminal, the first component terminal being coupled to the secondtransistor terminal. The switch includes a first switch terminal and asecond switch terminal, the first switch terminal being coupled to thesecond component terminal, the second switch. terminal being coupled tothe fifth transistor terminal. The first resistor includes a firstresistor terminal and a second resistor terminal, the first resistorbeing associated with a first resistance, the first resistor terminalbeing coupled to the second input terminal. The second resistor includesa third resistor terminal and a fourth resistor terminal, the secondresistor being associated with a second resistance, the third resistorterminal being coupled to the fourth input terminal. The firstresistance is larger than the second resistance in magnitude. The secondcomponent terminal is configured to output an output signal based on atleast information associated with the reference signal and the inputsignal. For example, the error amplifier is implemented according to atleast FIG. 7, FIG. 8, FIG. 9, FIG. 10, and/or FIG. 11.

According to another embodiment, a system controller for regulating apower conversion system includes a first controller terminal configuredto receive a first voltage associated with a first current, the firstcurrent being related to an input voltage of a power conversion system,a compensation component coupled to the first controller terminal andconfigured to, if the first voltage satisfies one or more firstconditions, generate a compensation current based on at leastinformation associated with the first current, and a second controllerterminal coupled to the compensation component and configured to providea compensation voltage based on at least information associated with thecompensation current, the compensation voltage being equal in magnitudeto the compensation current multiplied by a compensation resistance, thecompensation resistance being associated with a compensation resistor.The system controller further includes a current sensing componentconfigured to receive a second voltage and generate an output signal,the second voltage being equal to a sum of a third voltage and thecompensation voltage in magnitude, the third voltage being proportionalto a second current flowing through a primary winding of the powerconversion system, and an error amplifier configured to receive theoutput signal and a reference signal, generate an amplified signal basedon at least information associated with the output signal and thereference signal, and output the amplified signal to affect a switchassociated with the second current. For example, the system controlleris implemented according to at least FIG. 13.

According to yet another embodiment, a system controller for regulatinga power conversion system includes a first controller terminalconfigured to receive a feedback signal associated with an outputvoltage of a power conversion system, a compensation component coupledto the first controller terminal and configured to sample the feedbacksignal during a demagnetization process of the power conversion systemand generate a compensation current based on at least informationassociated with the sampled feedback signal, and a second controllerterminal coupled to the compensation component and configured to providea compensation voltage based on at least information associated with thecompensation current, the compensation voltage being equal in magnitudeto the compensation current multiplied by a compensation resistance, thecompensation resistance being associated with a compensation resistor.In addition, the system controller includes a current sensing componentconfigured to receive an input voltage and generate an output signal,the input voltage being equal to a sum of a first voltage and thecompensation voltage in magnitude, the first voltage being proportionalto a first current flowing through a primary winding of the powerconversion system, and an error amplifier configured to receive theoutput signal and a reference signal, generate an amplified signal basedon at least information associated with the output signal and thereference signal, and output the amplified signal to affect a switchassociated with the first current. For example, the system controller isimplemented according to at least FIG. 15, and/or FIG. 16.

According to yet another embodiment, a system controller for regulatinga power conversion system includes a first controller terminalconfigured to receive a feedback signal associated with an outputvoltage of a power conversion system, a compensation component coupledto the first controller terminal and configured to sample the feedbacksignal during a demagnetization process of the power conversion systemand generate a compensation current based on at least informationassociated with the sampled feedback signal, and an error amplifierincluding a first input terminal, a second input terminal, and an outputterminal coupled to the compensation component. The first input terminalis configured to receive an input voltage, the second input terminal isconfigured to receive a reference voltage, and the output terminal isconfigured to output a first output current related to a differencebetween the input voltage and the reference voltage in magnitude. Theerror amplifier and the compensation component are further configured togenerate a second output current equal to a difference between the firstoutput current and the compensation current in magnitude. For example,the system controller is implemented according to at least FIG. 19.

In one embodiment, a method for regulating a power conversion systemincludes receiving a first voltage associated with a first current, thefirst current being related to an input voltage of a power conversionsystem, generating, if the first voltage satisfies one or more firstconditions, a compensation current based on at least informationassociated with the first current, and providing an compensation voltagebased on at least information associated with the compensation current,the compensation voltage being equal in magnitude to the compensationcurrent multiplied by a compensation resistance, the compensationresistance being associated with a compensation resistor. The methodfurther includes receiving an input voltage, the input voltage beingequal to a sum of a second voltage and the compensation voltage inmagnitude, the second voltage being proportional to a second currentflowing through a primary winding of the power conversion system,generating an output signal based on at least information associatedwith the input voltage, and receiving the output signal and a referencesignal. In addition, the method includes generating an amplified signalbased on at least information associated with the output signal and thereference signal, and outputting the amplified signal in order to affecta switch associated with the second current. For example, the method isimplemented according to at least FIG. 13.

In another embodiment, a method for regulating a power conversion systemincludes receiving a feedback signal associated with an output voltageof a power conversion system, sampling the feedback signal during ademagnetization process of the power conversion system, and generating acompensation current based on at least information associated with thesampled feedback signal. The method further includes providing acompensation voltage based on at least information associated with thecompensation current, the compensation voltage being equal in magnitudeto the compensation current multiplied by a compensation resistance, thecompensation resistance being associated with a compensation resistor,receiving an input voltage, the input voltage being equal to a sum of afirst voltage and the compensation voltage in magnitude, the firstvoltage being proportional to a first current flowing through a primarywinding of the power conversion system, and generating an output signal.In addition, the method includes receiving the output signal and areference signal, generating an amplified signal based on at leastinformation associated with the output signal and the reference signal,and outputting the amplified signal to affect a switch associated withthe first current. For example, the method is implemented according toat least FIG. 15 and/or FIG. 16.

In yet another embodiment, a method for regulating a power conversionsystem includes receiving a feedback signal associated with an outputvoltage of a power conversion system, sampling the feedback signalduring a demagnetization process of the power conversion system, andgenerating a compensation current based on at least informationassociated with the sampled feedback signal. Additionally, the methodincludes receiving an input voltage at a first input terminal of anerror amplifier, and the error amplifier further includes a second inputterminal and an output terminal. Moreover, the method includes receivinga reference voltage at the second input terminal, generating a firstoutput current at the output terminal related to a difference betweenthe input voltage and the reference voltage in magnitude, and outputtinga second output current equal to a difference between the first outputcurrent and the compensation current in magnitude. For example, themethod is implemented according to at least FIG. 19.

For example, some or all components of various embodiments of thepresent invention each are, individually and/or in combination with atleast another component, implemented using one or more softwarecomponents, one or more hardware components, and/or one or morecombinations of software and hardware components. In another example,some or all components of various embodiments of the present inventioneach are, individually and/or in combination with at least anothercomponent, implemented in one or more circuits, such as one or moreanalog circuits and/or one or more digital circuits. In yet anotherexample, various embodiments and/or examples of the present inventioncan be combined.

Although specific embodiments of the present invention have beendescribed, it will be understood by those of skill in the art that thereare other embodiments that are equivalent to the described embodiments.Accordingly, it is to be understood that the invention is not to belimited by the specific illustrated embodiments, but only by the scopeof the appended claims.

What is claimed is:
 1. An error amplifier for a reference signal and aninput signal associated with a current of a power converter, the erroramplifier comprising: a first operational amplifier including a firstinput terminal, a second input terminal and a first output terminal, thefirst input terminal being configured to receive a reference signal, thefirst output terminal configured to output a first amplified signal to afirst transistor terminal of a first transistor, the first transistorfurther including a second transistor terminal and a third transistorterminal, and the second input terminal being coupled to the thirdtransistor terminal; a second operational amplifier including a thirdinput terminal, a fourth input terminal and a second output terminal,the third input terminal being configured to receive an input signalassociated with a first current flowing through a primary winding of apower converter, the second output terminal configured to output asecond amplified signal to a fourth transistor terminal of a secondtransistor, the second transistor further including a fifth transistorterminal and a sixth transistor terminal, and the fourth input terminalbeing coupled to the sixth transistor terminal; a current mirrorincluding a first mirror terminal and a second mirror terminal, thefirst mirror terminal being coupled to the second transistor terminal; aswitch including a first switch terminal and a second switch terminal,the first switch terminal being coupled to the second mirror terminal,the second switch terminal being coupled to the fifth transistorterminal; a first resistor including a first resistor terminal and asecond resistor terminal, the first resistor being associated with afirst resistance, the first resistor terminal being coupled to thesecond input terminal; and a second resistor including a third resistorterminal and a fourth resistor terminal, the second resistor beingassociated with a second resistance, the third resistor terminal beingcoupled to the fourth input terminal; wherein: the first resistance islarger than the second resistance in magnitude; and the error amplifieris configured to output a first output signal based at least in part onthe reference signal and the input signal.
 2. The error amplifier ofclaim 1 wherein the second operational amplifier is configured toreceive a clock signal, and change the second amplified signal at thesecond output terminal in response to the clock signal.
 3. The erroramplifier of claim 1 wherein: the error amplifier is coupled to aline-voltage compensator; and the line-voltage compensator is configuredto affect the input signal to keep an output current of the powerconverter approximately constant in response to an input voltagereceived by the power converter varying within a range.
 4. The erroramplifier of claim 3 wherein: the error amplifier is coupled to a signalgenerator; the line-voltage compensator is coupled to a first controllerterminal and a second controller terminal; the first controller terminalis configured to receive a first voltage associated with a secondcurrent, the second current being related to an input voltage of thepower converter; the line-voltage compensator is further configured to,in response to the first voltage satisfying one or more firstconditions, generate a compensation current based at least in part onthe second current; the second controller terminal is configured toprovide a compensation voltage based at least in part on thecompensation current, the compensation voltage being equal in magnitudeto the compensation current multiplied by a compensation resistance, thecompensation resistance being associated with a compensation resistor;the signal generator is configured to receive a second voltage andgenerate a second output signal, the second voltage being equal to a sumof a third voltage and the compensation voltage in magnitude, the thirdvoltage being proportional to the first current flowing through theprimary winding of the power converter; and the third input terminal isfurther configured to receive the second output signal.
 5. The erroramplifier of claim 1 wherein: the error amplifier is coupled to aload-signal compensator; and the load-signal compensator is configuredto affect the input signal to keep an output current of the powerconverter approximately constant if an output voltage of the powerconverter varies within a range.
 6. The error amplifier of claim 5wherein: the error amplifier is coupled to a signal generator; theload-signal compensator is coupled to a first controller terminal and asecond controller terminal; the first controller terminal is configuredto receive a feedback signal associated with an output voltage of thepower converter; the load-signal compensator is further configured tosample the feedback signal during a demagnetization process of the powerconverter and generate a compensation current based at least in part onthe sampled feedback signal; the second controller terminal isconfigured to provide a compensation voltage based at least in part onthe compensation current, the compensation voltage being equal inmagnitude to the compensation current multiplied by a compensationresistance, the compensation resistance being associated with acompensation resistor; the signal generator is configured to receive aninput voltage and generate a second output signal, the input voltagebeing equal to a sum of a first voltage and the compensation voltage inmagnitude, the first voltage being proportional to the first currentflowing through the primary winding of the power converter; and thethird input terminal is further configured to receive the second outputsignal.
 7. The error amplifier of claim 5 wherein: the load-signalcompensator is coupled to a first controller terminal and a secondcontroller terminal; the first controller terminal is configured toreceive a feedback signal associated with an output voltage of the powerconverter; the load-signal compensator is further configured to samplethe feedback signal during a demagnetization process of the powerconverter and generate a compensation signal based at least in part onthe sampled feedback signal; the error amplifier is coupled to a thirdcontroller terminal; the third controller terminal is configured toprovide the second output terminal; a second output signal is equal to adifference between a third amplified signal and the compensation signalin magnitude; and the third amplified signal is related to a differencebetween the input signal and the reference signal in magnitude.
 8. Theerror amplifier of claim 1 wherein the switch is configured to receive ademagnetization signal and be closed or opened in response to thedemagnetization signal.
 9. The error amplifier of claim 1 wherein theswitch is configured to be closed during a demagnetization periodassociated with the power converter.
 10. The error amplifier of claim 1wherein the switch is configured to be opened before and after ademagnetization period associated with the power converter.
 11. Theerror amplifier of claim 8 wherein the error amplifier is furtherconfigured to output the first output signal based at least in part onthe reference signal, the input signal, and the demagnetization signal.12. The error amplifier of claim 1 wherein: the current mirror includes:a third transistor including a seventh transistor terminal, an eighthtransistor terminal and a ninth transistor terminal; and a fourthtransistor including a tenth transistor terminal, an eleventh transistorterminal and a twelfth transistor terminal; wherein: the seventhtransistor terminal is coupled to the ninth transistor terminal and thetenth transistor terminal; the eighth transistor terminal is coupled tothe eleventh transistor terminal; and the ninth transistor terminal iscoupled to the second transistor terminal.
 13. The error amplifier ofclaim 12 wherein the twelfth transistor terminal is coupled to the firstswitch terminal.
 14. The error amplifier of claim 12 wherein: thecurrent mirror further includes: a fifth transistor including athirteenth transistor terminal, a fourteenth transistor terminal and afifteenth transistor terminal; and a third operational amplifierincluding a fifth input terminal, a sixth input terminal and a thirdoutput terminal; wherein: the thirteenth transistor terminal is coupledto the third output terminal; the fourteenth transistor terminal iscoupled to the twelfth transistor terminal and the fifth input terminal;the fifteenth transistor terminal is coupled to the first switchterminal; and the sixth input terminal is coupled to the ninthtransistor terminal and the second transistor terminal.